Calibrated temperature measurement system

ABSTRACT

In accordance with some embodiments of the present disclosure, a calibrated temperature measurement system comprises a resistor, a thermistor, a resistance-to-current converter configured to generate a current signal based on a resistance, and an analog-to-digital converter (ADC) configured to receive a first current signal based on the resistor, convert the first current signal into a first digital signal, receive a second current signal based on the thermistor, and convert the second current signal into a second digital signal. A memory may comprise resistor-characterization information. A calculation stage communicatively coupled to an ADC output may be configured to determine a first digital value based on the first digital signal, determine a second digital value based on the second digital signal, calculate a resistance ratio based on the first digital value and the second digital value, and determine a temperature output value based on the resistance ratio and the resistor-characterization information.

TECHNICAL FIELD

The present disclosure relates generally to electrical circuits and,more particularly, to temperature-measurement and voltage-measurementcircuits.

BACKGROUND

The temperature at which a circuit is operating is a key performanceconsideration in many applications. For example, the performance ofvarious semiconductor devices may vary widely across hot and coldtemperatures. Thus, semiconductor manufacturers often guarantee circuitperformance only for a predetermined temperature range. Becausetemperature can be a key performance consideration, some applicationsactively measure and monitor temperature. Devices known as thermistorsare known to have a resistances that vary over temperature. Accordingly,some applications measure the resistance of a thermistor in order totrack the approximate temperature at which a circuit is operating.

SUMMARY

A calibrated temperature measurement system is disclosed. In accordancewith some embodiments of the present disclosure, a calibratedtemperature measurement system comprises a resistor, a thermistor, aresistance-to-current converter configured to generate a current signalbased on a resistance, and an analog-to-digital converter (ADC)configured to receive a first current signal based on the resistor,convert the first current signal into a first digital signal, receive asecond current signal based on the thermistor, and convert the secondcurrent signal into a second digital signal. The system may furthercomprise a memory comprising resistor-characterization information and acalculation stage communicatively coupled to an ADC output andconfigured to determine a first digital value based on the first digitalsignal, determine a second digital value based on the second digitalsignal, calculate a resistance ratio based on the first digital valueand the second digital value, and determine a temperature output valuebased on the resistance ratio and the resistor-characterizationinformation.

In accordance with another embodiment of the present disclosure, acalibrated temperature measurement system may comprise a resistor, aresistance-to-current converter configured to generate a current signalbased on a resistance, and an analog-to-digital converter (ADC)configured to receive a first current signal, the first current signalbased on the resistor, convert the first current signal into a firstdigital signal, receive a second current signal, and convert the secondcurrent signal into a second digital signal. The system may alsocomprise a memory comprising resistor-characterization information and acalculation stage communicatively coupled to an ADC output andconfigured to determine a first digital value based on the first digitalsignal, determine a second digital value based on the second digitalsignal, calculate a resistance ratio based on the first digital valueand the second digital value, and determine a temperature output valuebased on the resistance ratio and the resistor-characterizationinformation.

In accordance with another embodiment of the present disclosure, amethod for characterizing a temperature measurement system comprisesgenerating a first current signal based on a reference resistor,converting the first current signal into a first digital signal,determining a first digital value corresponding to a resistance of thereference resistor based on the first digital signal, generating asecond current signal based on a test resistor, converting the secondcurrent signal into a second digital signal, determining a seconddigital value corresponding to a resistance of the test resistor basedon the second digital signal, and storing reference-resistorcharacterization information in a memory.

In accordance with yet another embodiment of the present disclosure, amethod for measuring temperature, comprises generating a first currentsignal based on a resistor, converting the first current signal into afirst digital signal, determining a first digital value corresponding toa resistance of the resistor based on the first digital signal,generating a second current signal based on a thermistor, converting thesecond current signal into a second digital signal, determining a seconddigital value corresponding to a resistance of the thermistor based onthe second digital signal, calculating a resistance ratio based on thefirst digital value and the second digital value, and determining atemperature output value based on the resistance ratio and aresistor-characterization ratio.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory and arenot restrictive of the invention, as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete and thorough understanding of the present embodimentsand advantages thereof may be acquired by referring to the followingdescription taken in conjunction with the accompanying drawings, inwhich like reference numbers indicate like features, and wherein:

FIG. 1A illustrates a schematic diagram of a current-mode buffer, inaccordance with the teachings of the present disclosure;

FIG. 1B illustrates a schematic diagram of a current-mode buffer, inaccordance with the teachings of the present disclosure;

FIG. 2 depicts a flow chart of an example method for measuring theresistances of a resistor and a thermistor, in accordance with theteachings of the present disclosure;

FIG. 3 illustrates a block diagram depicting a temperature measurementsystem, in accordance with the teachings of the present disclosure;

FIG. 4 depicts a flow chart of an example method for measuringtemperature, in accordance with the teachings of the present disclosure;

FIG. 5 illustrates a block diagram depicting a calibrated temperaturemeasurement system, in accordance with the teachings of the presentdisclosure;

FIG. 6 illustrates a block diagram of a look-up map, in accordance withthe teachings of the present disclosure;

FIG. 7 depicts a flow chart of an example method for calibrating atemperature measurement system, in accordance with the teachings of thepresent disclosure;

FIG. 8 depicts a flow chart of an example method for measuringtemperature, in accordance with the teachings of the present disclosure;

FIG. 9 illustrates a schematic diagram of a current-mode buffer, inaccordance with the teachings of the present disclosure;

FIG. 10 illustrates a schematic diagram of an amplifier, in accordancewith the teachings of the present disclosure; and

FIG. 11 illustrates a block diagram depicting a voltage measurementsystem, in accordance with the teachings of the present disclosure.

DETAILED DESCRIPTION

FIG. 1A illustrates a schematic diagram of a current-mode buffer 100, inaccordance with the teachings of the present disclosure. Buffer 100 maybe an input stage for a system that, as described in greater detail withreference to FIG. 3, may be configured to measure temperature bymeasuring the value of a temperature-dependant resistance.

Buffer 100 may include a common-mode voltage reference (V_(CM)), N:1multiplexors (MUX) 111, an amplifier 120, p-type metal-oxidesemiconductor field-effect transistors (PMOS) 130, 132, 134, n-typemetal-oxide semiconductor field-effect transistors (NMOS) 140, 142, 144,as well as switch 112. Buffer 100 may be configured to sense aresistance coupled to one of multiple inputs of buffer 100 (e.g.,V_(IN1) and V_(IN2)), and to output a current based on the value of theresistance. The output current may be a differential output current,I_(OUT) ⁺ and I_(OUT) ⁻. In some embodiments, the output current may beinversely proportional to, or otherwise based on the sensed resistance.Buffer 100 may be configured to sense the resistance of any suitabledevice coupled to one of its inputs. For example, buffer 100 may beconfigured to sense the resistance of a thermistor, a resistor, a diode,a diode-connected transistor, or any other device with a resistivecharacteristic. Such resistive devices may off-chip devices or on-chipdevices located on the same semiconductor chip as buffer 100. For thepurposes of the present disclosure, a thermistor may be any device whoseresistance varies in a known manner with temperature. Accordingly, thecurrent through a thermistor may vary across a range of temperatureswhen a given voltage is applied across the thermistor. Some commerciallyavailable thermistors may be case grounded, and thus may have oneterminal coupled to ground (GND). Accordingly, some embodiments ofbuffer 100 may be configured to sense the resistance of a device coupledbetween an input of buffer 100 and GND.

As described above, buffer 100 may be configured to operate as an inputstage for a temperature measurement system. In some embodiments, such atemperature measurement system may be configured to determine thetemperature based on the measured resistance of thermistor 105. In someembodiments, such a temperature measurement system may be configured todetermine the temperature based on a ratio including both the resistanceof thermistor 105 and the resistance of reference resistor 106.Accordingly, buffer 100 may be configured to sense the resistance ofdifferent devices (e.g., thermistor 105 and reference resistor 106) atdifferent times. For example, thermistor 105 may be coupled to V_(IN1)and reference resistor 106 may be coupled to V_(IN2). During a firstperiod of time during which the resistance of thermistor 105 may besensed and/or measured, MUX 111 b may couple V_(IN1) to an input ofamplifier 120, and MUX 111 a may couple a feedback path (e.g., a currentpath including PMOS 130) to V_(IN1). Likewise, during a second period oftime during which the resistance of reference resistor 106 may be sensedand/or measured, MUX 111 b may couple V_(IN2) to an input of amplifier120, and MUX 111 a may couple a feedback path (e.g., a current pathincluding PMOS 130) to V_(IN2).

Though MUX 111 a and MUX 111 b may be illustrated as 2:1 multiplexors,MUX 111 a and MUX 111 b may be “N:1” multiplexors with any suitable “N”number of inputs. Accordingly, MUX 111 a and MUX 111 b may accommodateany suitable number of inputs (e.g., three, four, or more) for anysuitable number of resistive devices to be sensed and/or measured bybuffer 100.

PMOS 130 and amplifier 120 may be configured to generate a current basedon the resistance coupled to amplifier 120. As described above, when thepositive input terminal of amplifier 120 is coupled by MUX 111 b toV_(IN1), the feedback path including PMOS 130 may also be coupled toV_(IN1). At this time, amplifier 120 may compare the voltage at V_(IN1)to a reference voltage (e.g., common-mode reference V_(CM)). The outputof amplifier 120 may then drive PMOS 130 via feedback node 125 togenerate a feedback current, I_(FB). The feedback current may in turncreate a voltage drop across thermistor 105 such that the voltage atV_(IN1) is equivalent to the voltage of V_(CM). Though the voltage atV_(IN1) may be equivalent to the voltage of V_(CM), there may be someoffset due to an offset of amplifier 120 that may be the result of oneor more non-idealities (e.g., semiconductor device mismatch, processingerrors, and/or a limited voltage gain of amplifier 120). The feedbackcurrent I_(FB) during the sensing and/or measurement of thermistor 105may be represented by the following equation: I_(FB)=V_(IN1)/R_(TH),where R_(TH) may be the resistance of thermistor 105. Because thevoltage at V_(IN1) may be forced by amplifier 120 to be equivalent toV_(CM), the feedback current I_(FB) may also be expressed asI_(FB)=V_(CM)/R_(TH). Buffer 100 may operate in a similar manner whenthe resistance of reference resistor 106 coupled to V_(IN2) is sensedand/or measured. For such measurements, the feedback current I_(FB) maybe expressed as I_(FB)=V_(CM)/R_(REF), where R_(REF) may be theresistance of reference resistor 106.

MUX 111 a and MUX 111 b may include components, such as pass-gatetransistors (not expressly shown), that may have a resistance.Accordingly, any current flowing through MUXs 111 may cause a voltagedrop. Because the resistances of such devices internal to MUXs 111 maybe affected by various parameters (e.g., semiconductor process variationand/or temperature) any voltage drop across MUX 111 may vary. To avoiderrors caused by such varying voltage drops, the inputs of amplifier 120may be configured to not draw any significant amount of current.Accordingly, buffer 100 may sense a resistance coupled to V_(IN1) and/orV_(IN2) without a significant amount of current flowing through MUX 111b. Thus, any voltage drop and/or error associated with such currentthrough MUX 111 b may be avoided. Because buffer 100 may be configuredto sense the value of resistance coupled to V_(IN1) and/or V_(IN2)without drawing any current from V_(IN1) and/or V_(IN2) to the input ofamplifier 120, buffer 100 may be referred to herein as a “high-impedancebuffer.”

As described above, the value of the feedback current, I_(FB), may basedon value of the resistance coupled to V_(IN1) or V_(IN2). Thus, acurrent that may be equivalent to I_(FB), proportional to I_(FB), orotherwise based on I_(FB), may be utilized to measure the value of theresistance coupled to V_(IN1) or V_(IN2). In some embodiments, PMOS 132may be configured to mirror PMOS 130, and thereby may provide asingle-ended output current, I_(OUT), that may be proportional toI_(FB). Like PMOS 130, PMOS 132 may have a source coupled to VDD and agate coupled to feedback node 125.

In order to scale I_(OUT) to a desired level for a given I_(FB), thesize of PMOS 132 may be adjusted to any suitable size that may be largeror smaller than the size of PMOS 130. For the purposes of the presentdisclosure, the “size” of a PMOS or an NMOS transistor may refer to thewidth-to-length ratio of the transistor. In some embodiments, the sizeof PMOS 132 may be configured to be double the size of PMOS 130, andthus the value of I_(OUT) may be double the value of I_(FB). Likewise,in some embodiments, the size of PMOS 132 may be configured to be halfthe size of PMOS 130, and thus the value of I_(OUT) may be half thevalue of I_(FB). For the purposes of the present invention, the ratioI_(OUT) to I_(FB) may be referred to as the “gain” of buffer 100.

In some embodiments, the gain of buffer 100 may be dynamically adjusted.For example, PMOS 132 may have an adjustable size. To implement anadjustable size, PMOS 132 may include any suitable number of individualPMOS devices that may be selectively included in the operation of PMOS132. Each individually selected PMOS device may contribute to theoverall size of PMOS 132. Accordingly, the selection of more individualPMOS devices may result in a larger effective size for PMOS 132, and theselection of less individual PMOS devices may result in a smallereffective size for PMOS 132. In some embodiments, the selection andde-selection of such individual devices may be implemented by thecoupling of one or more of the gate, source, and/or drain terminals ofeach individual device to the respective gate, source, and/or drainterminals of PMOS 132 via a switch, pass-gate, or any other suitabledevice configured to selectively couple two terminals together.

Buffer 100 may be configured to output I_(OUT) to either the positiveoutput node 154 or the negative output node 152. For example, the outputstage of buffer 100 may include switch 112. During a first polaritystate, switch 112 may route the single-ended I_(OUT) to positive outputnode 154, and during a second polarity state, switch 112 may route thesingle-ended I_(OUT) to negative output node 152. The alternatingpolarities of the output of buffer 100 may be used by a systemincorporating buffer 100 to correct for offsets and/or other signalerrors caused by buffer 100 due to semiconductor device mismatch,semiconductor processing defects, or other types of non-idealities. Forexample, buffer 100 may incur an offset due to, for example, a positiveoffset at the input of amplifier 120 during a first measurement ofthermistor 105. During the first measurement of thermistor 105, thesingle-ended I_(OUT) may be routed to positive output terminal 154, andthus the offset may be output as positive offset. The polarity state ofbuffer 100 may then be alternated by alternating the state of switch112. During a second measurement of thermistor 105, the single-endedI_(OUT) may be routed to negative output terminal 152. A similar offsetmay be incurred during the second measurement. But because I_(OUT) maybe routed to negative output terminal 152, the offset may be output as anegative offset. For the purposes of the present disclosure, an outputcurrent based on a resistive device (e.g., thermistor 105) while buffer100 is in a second polarity state may be referred to as being“complementary” of an output current based on that same resistive devicewhile buffer 100 is in a first polarity state. As described in furtherdetail below with reference to FIG. 3, a system receiving the two outputcurrents representing the first and second measurements of thermistor105 may combine the two measurements such that certain offsets generatedduring the first and second polarity states cancel out.

In some embodiments, buffer 100 may be configured to convert thesingle-ended I_(OUT) into a differential output current, I_(OUT) ⁺ andI_(OUT) ⁻. In order to convert a singled-ended current signal such asI_(OUT) into a differential current signal, buffer 100 may be configuredto source I_(OUT) to one of the two output nodes 152 and 154 whilesinking one-half I_(OUT) from both output nodes 152 and 154. Such sourceand sink currents may result in a positive current of approximatelyone-half I_(OUT) at one of the two output nodes 152 and 154, and anegative current of approximately one-half I_(OUT) at the other of thetwo output nodes 152 and 154. Accordingly, a differential outputcurrent, I_(OUT) ⁺ and I_(OUT) ⁻, may be provided at output nodes 152and 154 with a differential value that may be approximately equivalentto the single-ended value of I_(OUT).

In some embodiments, buffer 100 may include PMOS 134. Like PMOS 132,PMOS 134 may have a gate coupled to feedback node 125 and a sourcecoupled to VDD. PMOS 134 may be configured to match PMOS 132, but with asize that may be approximately one-half of the size of PMOS 132.Accordingly, the current provided by PMOS 134 may be approximatelyone-half of I_(OUT). NMOS 140, NMOS 142, and NMOS 144 may in turn beconfigured to mirror the one-half-I_(OUT) current of PMOS 134. Forexample, NMOS 140 may have a source coupled to ground (“GND”), and agate and a drain coupled to the drain of PMOS 134. As such, NMOS 140 maygenerate a gate bias and may sink the one-half-I_(OUT) current providedby PMOS 134. NMOS 142 and NMOS 144 may each include a source coupled toGND and a gate coupled to the gate of NMOS 140. NMOS 142 and NMOS 144may be configured to match each other and may have approximately thesame size. Moreover, NMOS 142 and NMOS 144 may configured to match NMOS140 and may have approximately the same size as NMOS 140. Accordingly,NMOS 142 and NMOS 144 may each be configured to sink a current that maybe approximately equal to one-half I_(OUT). Thus, NMOS 142 may sink acurrent of approximately one-half I_(OUT) from output node 152, and NMOS144 may sink a current of approximately one-half I_(OUT) from outputnode 154. For the purposes of the present disclosure, NMOS 142 and NMOS144 may be referred to either as a current sink or as a current source.Though the common mode of the output of buffer may be set by sourcingI_(OUT) to one of the two output nodes 152 and 154 while also sinkingone-half I_(OUT) currents from both output nodes 152 and 154, the commonmode of the output of buffer 100 may be established in any suitablemanner. For example, I_(OUT) could be sunk from one of the two outputnodes 152 and 154, while one-half I_(OUT) currents are sourced to bothof the two output nodes 152 and 154.

Though the halving of I_(OUT) is described above as a result of the PMOS134 have one-half the size of PMOS 132, the halving of I_(OUT) may beimplemented in any suitable manner. For example, in some embodiments,PMOS 134 may have the same size as PMOS 132, and thus may provide acurrent equivalent to I_(OUT) to NMOS 140. In such embodiments, NMOS 142and NMOS 144 may be half the size of NMOS 140.

In some embodiments, I_(OUT) may be mirrored from I_(FB) without beingdirectly driven by amplifier 120. FIG. 1B illustrates a schematicdiagram of a current-mode buffer 102, in accordance with the teachingsof the present disclosure. As illustrated in FIG. 1B, amplifier 120 maybe configured to drive NMOS 135. NMOS 135 may have a gate coupled to theoutput of amplifier 120, a source coupled to MUX 111 a, and a draincoupled to the gate and drain of a diode-connected device such as PMOS136. PMOS 136 may have source coupled to VDD, and, as described above, agate and a drain coupled to together and to the drain of NMOS 135.Because PMOS 136 may be in the same current path as the feedbacktransistor NMOS 135, the diode-connected PMOS 136 may self-bias with thefeedback current I_(FB). PMOS 138 and PMOS 137 may each have a gatecoupled to the gate of PMOS 136, and accordingly may mirror PMOS 136.For example, PMOS 138 may mirror PMOS 136 to generate I_(OUT) in asimilar manner as PMOS 132 mirrors PMOS 130 in FIG. 1A. Likewise, PMOS137 may mirror PMOS 136 to generate a one-half I_(OUT) in a similarmanner as PMOS 134 mirrors PMOS 130 in FIG. 1A.

As described above, the ability of buffer 100 to alternate polaritiesduring two measurements of a device such as thermistor 105 may allow asystem implementing buffer 100 to effectively cancel any offset incurredduring the two measurements. Because the architecture of buffer 100 mayallow for offsets to be canceled out at a later stage, buffer 100 may bedesigned with relaxed requirements for various parameters that maycontribute to offset. For example, the dimensions of matched transistorsin amplifier 120 (e.g., a differential pair and/or a current mirror) mayaffect how closely those matched transistors actually match. Typically,transistors with smaller dimensions (e.g., channel width and channellength for NMOS or PMOS devices) may be more susceptible tosemiconductor processing defects or mismatch than transistors withlarger dimensions. Such semiconductor processing defects or mismatch mayaffect, for example, a differential pair of transistors (not expresslyshown) that may form the input to amplifier 120. The result of suchmismatch may be an offset across the inputs of amplifier 120, which mayin turn result in an offset in the output current. However, because anymismatch-induced offset may be cancelled out at a later stage, matchedtransistors within amplifier 120 may be implemented with nominaldimensions. Thus, a significant amount of semiconductor area may besaved as compared to matched transistors that are implemented with largedimensions in order to minimize mismatch and offset.

FIG. 2 depicts a flow chart of an example method 200 for measuring theresistances of a resistor and a thermistor, in accordance with theteachings of the present disclosure.

At step 202, a resistor may be coupled to an amplifier input of abuffer. For example, reference resistor 106 may be selectively coupledby MUX 111 b to the positive input of amplifier 120, and the feedbackpath may be coupled to reference resistor 106 by MUX 111 a.

At step 204, the buffer may be set to a first polarity state. Forexample, switch 112 may be set to route the single-ended I_(OUT) to thepositive output node 154, while NMOS 144 and NMOS 142 each sink acurrent of one-half I_(OUT) from positive output node 154 and negativeoutput node 152 respectively.

At step 206, a first output current may be generated based on theresistor. For example, amplifier 120 may drive feedback node 125 to avoltage that causes a transistor in the feedback path (e.g., PMOS 130)to generate a feedback current that may be sufficient to force thevoltage across reference resistor 106 to be equivalent to V_(CM). Inaddition, PMOS 132 may mirror the feedback current of PMOS 130. In someembodiments, the single-ended signal current from PMOS 132 may be outputas a single-ended output current. In some embodiments, the single-endedsignal current from PMOS 132 may be converted into a differential outputcurrent by the one-half-I_(OUT) current sinks formed by NMOS 142 andNMOS 144.

At step 208, the buffer may be set to a second polarity state. Forexample, switch 112 may be set to route the single-ended I_(OUT) to thenegative output node 152, while NMOS 144 and NMOS 142 each sink acurrent of one-half I_(OUT) from positive output node 154 and negativeoutput node 152 respectively.

At step 210, a first complementary output current may be generated basedon a resistance of the resistor. Though the polarity of buffer 100 maybe set to a second polarity state during step 210, a first complementaryoutput current based on reference resistor 106 may be generated bybuffer 100 in a manner similar to that of step 206.

At step 212, a thermistor may be coupled to an amplifier input of thebuffer. For example, thermistor 105 may be selectively coupled by MUX111 b to the positive input of amplifier 120, and the feedback path maybe coupled to thermistor 105 by MUX 111 a.

During steps 214 through 220, multiple output currents based on athermistor may be generated in a similar manner as the multiple outputcurrents based on the resistor generated in steps 204 through 210.

At step 214, buffer 100 may be re-set to the first polarity statedescribed in step 204.

At step 216, a second output current may be generated based on aresistance of the thermistor. For example, amplifier 120 may drivefeedback node 125 to a voltage that causes a transistor in the feedbackpath (e.g., PMOS 130) to generate a feedback current that may besufficient to force the voltage across thermistor 105 to be equivalentto V_(CM). In addition, PMOS 132 may mirror the feedback current of PMOS130. In some embodiments, the single-ended signal current from PMOS 132may be output as a single-ended output current. In some embodiments, thesingle-ended signal current from PMOS 132 may be converted into adifferential output current by the one-half-I_(OUT) current sinks formedby NMOS 142 and NMOS 144.

At step 218, buffer 100 may be re-set to the second polarity statedescribed in step 208. With buffer 100 in the second polarity state, asecond complementary output current may be generated at step 220 basedon a resistance of the thermistor. Though the polarity of buffer 100 maybe set to a second polarity state during step 220, the secondcomplementary output current may be generated by buffer 100 in anotherwise similar manner to that described above for step 216.

Although FIG. 2 discloses a particular number of steps to be taken withrespect to method 200, method 200 may be executed with greater or lessersteps than those depicted in FIG. 2. For example, method 200 may beexecuted with only steps 202, 206, 212, and 216. In addition, althoughFIG. 2 discloses a certain order of steps to be taken with respect tomethod 200, the steps included in method 200 may be completed in anysuitable order. For example, steps 212 through 220 may be performedprior to steps 202 through 210.

FIG. 3 illustrates a block diagram depicting a temperature measurementsystem 300, in accordance with the teachings of the present disclosure.Temperature measurement system 300 may include thermistor 105, referenceresistor 106, an analog input stage 305, and a digital calculation stage315. In some embodiments, analog input stage 305 and digital calculationstage 315 may be included on a single semiconductor chip, and thermistor105 and reference resistor 106 may be external components. For suchembodiments, thermistor 105 may be coupled to analog input stage 305 viapin 301, and reference resistor 106 may be coupled to analog input stage305 via pin 302.

Analog input stage 305 may include buffer 100 and analog-to-digitalconverter (ADC) 310. As described above with reference to FIG. 1, buffer100 may be configured to output a differential current signal that maybe inversely proportional to, or otherwise based on, the value of aresistance coupled to buffer 100. Moreover, buffer 100 may be configuredto alternate the polarity state of the output of buffer 100. Forexample, during a first measurement of reference resistor 106, buffer100 may output a current to its positive output node 154 which may becoupled to a positive input of ADC 310. Likewise, during a secondmeasurement of reference resistor 106, buffer 100 may output a currentto its negative output node 152, which may be coupled to a negativeinput of ADC 310. In some embodiments, analog input stage 305 mayinclude a resistance-to-current converter other than buffer 100. In suchembodiments, analog input stage 305 may include an alternating-polaritydevice, which may alternate the routing of the output of theresistance-to-current converter to the positive input of ADC 310 and thenegative input of ADC 310 in a similar manner as switch 112 alternatesthe output current of buffer 100. Moreover, in such embodiments, analoginput stage 305 may include a common-mode circuit which may set and/orcontrol the common mode of the current signal received by ADC 310.

ADC 310 may be implemented with any suitable type of ADC that may beconfigured to convert an analog current signal into a digital signal.For example, ADC 310 may be a sigma-delta ADC that may be configured toserially output a stream of digital bits that may represent the valuethe differential current signal, I_(OUT) ⁺ and I_(OUT) ⁻. Embodiments ofADC 310 that are implemented as a sigma-delta ADC, or any other suitabletype of ADC, may include an suitable number of integration stages. Insome embodiments, such integration stages may include a continuous-timeintegrator 312. The continuous-time operation of the one or morecontinuous-time integrators 312 may allow ADC 310 to operate in alow-noise manner. For example, continuous-time integrator 312 maygenerate significantly less noise than a switched-capacitor integrator.Such low-noise operation may allow ADC 310 to be located on the samesemiconductor chip as noise-sensitive circuits (e.g., a wirelesstransceiver). Such integration may reduce part count and associatedcosts in various applications by allowing more circuits to be integratedon a single semiconductor chip.

Digital calculation stage 315 may include digital logic configured toreceive and process one or more streams of digital bits from ADC 310 andto calculate a temperature. Digital calculation stage 315 may include ade-multiplexer (DEMUX) 330, one or more decimators 340, one or moreoffset cancellers 350, an adder 360, a divider 370, and a look-up map380. Digital calculation stage 315 may include logic implemented in anysuitable manner. For example, the logic of digital calculation stage 315may be implemented in an application-specific integrated circuit (ASIC),in a field-programmable gate array (FPGA), in program instructionsstored in a memory and configured to be executed by a multi-purposeprocessor, or any suitable combination thereof.

As described in further detail below, digital calculation stage 315 maybe configured to: (i) convert the stream of bits from ADC 310representing, at different times, the resistances of thermistor 105 andreference resistor 106 into multi-bit digital values (e.g., I_(TH) andI_(REF)); (ii) calculate a resistance ratio based on the two digitalvalues; and (iii) determine a temperature based on the calculatedresistance ratio.

In some embodiments, the input of digital calculation stage 315 may becoupled to the input of DEMUX 330. In some embodiments, DEMUX 330 may bea one-to-four demultiplexer and may be configured to couple the input ofdigital calculation stage 315 to one of four decimators 340 at a time.Each of the four decimators 340 a-d may be configured to receive thestream of digital bits from ADC 310 during one of four measurements(e.g., measurements M1 through M4). For example, DEMUX 330 may routemeasurement M1 to decimator 340 a, measurement M2 to decimator 340 b,measurement M3 to decimator 340 c, and measurement M4 to decimator 340d. Measurements M1 and M2 may include complementary measurements ofreference resistor 106. For example, measurement M1 may include thestream of bits from ADC 310 during a first period of time when buffer100 senses the resistance of reference resistor 106 while in a firstpolarity state. In addition, measurement M2 may include the stream ofbits from ADC 310 during a second period of time when buffer 100 sensesthe resistance of reference resistor 106 while in a second polaritystate. Similarly, measurements M3 and M4 may include complementarymeasurements for thermistor 105. For example, measurement M3 may includethe stream of bits from ADC 310 during a third period of time whenbuffer 100 senses the resistance of thermistor 105 while in a firstpolarity state. In addition, measurement M4 may include the stream ofbits from ADC 310 during a fourth period of time when buffer 100 sensesthe resistance of thermistor 105 while in a second polarity state.

Each decimator 340 may be configured to convert a stream of digital bitsfrom ADC 310 into a single multi-bit value. Decimator 340 may seriallyreceive any suitable number of bits and may output a single multi-bitvalue. For example, decimator 340 may serially receive sixty-fourconsecutive bits from ADC 310 and may output a single multi-bit valuecorresponding to the number of the sixty-four input bits that were setto logical one. In some embodiments, upon receiving a logical one,decimator 340 may add one to its output value. Likewise, upon receivinga logical zero, decimator 340 may subtract one from its output value.Accordingly, for the sixty-four consecutive input bits, decimator 340may have a minimum output value of negative sixty-four, and a maximumoutput value of positive sixty-four. Though the above example ofdecimator 340 refers to serially receiving sixty-four consecutive inputbits, decimator 340 may be configured to receive any suitable number ofbits in order to output a multi-bit value of suitable accuracy. Thebit-size of decimator 340 may depend on multiple factors including, butnot limited to, the desired resolution of the multi-bit output and thedesired signal range. For example, decimator 340 may be configured toconvert a large number of serially received bits in order to provide alarge enough signal range that may avoid saturation when the gain ofbuffer 100 is dynamically increased or decreased, as described abovewith reference to FIG. 1. Moreover, decimator 340 may be any suitablenumber of order decimator. For example, decimator 340 may be afirst-order decimator, a fourth-order decimator, or any other suitablenumber of order decimator.

Offset canceller 350 a may be configured to receive the multi-bit valuesfrom decimators 340 a and 340 b representing measurements M1 and M2, andto output a multi-bit digital value, I_(REF), which may represent thecurrent of reference resistor 106 during the measurements of referenceresistor 106. As described above, measurement M1 may have been performedon reference resistor 106 with buffer 100 set to a first polaritysetting, and measurement M2 may have been performed on referenceresistor 106 with buffer 100 set to a second polarity setting.Accordingly, offsets incurred during measurement M1 may correspond toequivalent offsets incurred during measurement M2. Such equivalentoffsets may be cancelled out by any suitable technique. For example, asshown by the equations one through three, offset canceller 350 a maysubtract the value received from decimator 340 b from the value receivedfrom decimator 340 a in order to cancel offset current.

The current-mode measurement of reference resistor 106 with buffer 100in a first polarity state may be represented as:

M1=(V _(CM) ′/R _(REF))+I _(OFF)  (Eq. One)

where V_(CM)′ is the common-mode reference voltage plus the offset ofamplifier 120 in buffer 100, R_(REF) is the resistance of referenceresistor 106, and I_(OFF) is the offset current incurred in or at theinput of ADC 310 (e.g., input offset current of ADC 310 and/or outputoffset current of buffer 100 caused by mismatch of NMOS 142 and NMOS144). Likewise the current-mode measurement of reference resistor 106with buffer 100 in a second polarity state may be represented as:

M2=−(V _(CM) ′/R _(REF))+I _(OFF).  (Eq. Two)

Subtracting M2 from M1 may accordingly result in the following:

I _(REF) =M1−M2=2*V _(CM) ′/R _(REF).  (Eq. Three)

As shown below, the value of I_(REF) may be further combined with thevalue of I_(TH) to further cancel the offset of amplifier 120represented in the value of V_(CM)′.

Offset canceller 350 b may be configured to operate in a similar manneras offset canceller 350 a. Offset canceller 350 b may be configured toreceive the multi-bit values from decimators 340 c and 340 drepresenting measurements M3 and M4, and to output a multi-bit digitalvalue, I_(TH), which may represent the current of thermistor 105 duringthe measurements of thermistor 105. As described above, measurement M3may have been performed on thermistor 105 with buffer 100 set to a firstpolarity setting, and measurement M4 may have been performed onthermistor 105 with buffer 100 set to a second polarity setting.Accordingly, offset incurred during measurement M3 may correspond to anequivalent offset incurred during measurement M4. Such equivalentoffsets may be cancelled out by any suitable technique. For example, asshown by the equations four through six, offset canceller 350 b maysubtract the value received from decimator 340 d from the value receivedfrom decimator 340 c in order to cancel offset current.

The current-mode measurement of thermistor 105 with buffer 100 in afirst polarity state may represented as:

M3=(V _(CM) ′/R _(TH))+I _(OFF)  (Eq. Four)

where V_(CM)′ is the common-mode reference voltage plus the offset ofamplifier 120 in buffer 100, R_(TH) is the resistance of thermistor 105,and I_(OFF) is the offset current of ADC 310. Likewise the current-modemeasurement of thermistor 105 with buffer 100 in a second polarity statemay be represented as:

M4=(−V _(CM) ′/R _(TH))+I _(OFF).  (Eq. Five)

Subtracting M4 from M3 may accordingly result in the following:

I _(TH) =M3−M4=2*V _(CM) ′/R _(TH).  (Eq. Six)

As shown below, the value of I_(TH) may be further combined with thevalue of I_(REF) to further cancel the offset of amplifier 120represented in the value of V_(CM)′.

After I_(TH) and I_(REF) are determined, I_(TH) and I_(REF) may becombined in a ratio. As shown by equations eight and nine, such acurrent ratio may be equivalent to a resistance ratio including therespective resistances of reference resistor 106 and thermistor 105. Forexample, adder 360 may add I_(TH) and I_(REF). Divider 370 may thendivide I_(REF) by the output of adder 360 (i.e., the sum of I_(TH) andI_(REF)). Substituting equation three and equation six for the values ofI_(REF) and I_(TH) results in the following:

I _(REF)/(I _(REF) +I _(TH))=(2*V _(CM) ′/R _(REF))/((2*V _(CM) ′/R_(REF))+(2*V _(CM) ′/R _(TH))).  (Eq. Seven)

In such a ratio, the factor of two, and the value of V_(CM)′ (whichincludes the offset of amplifier 120 in buffer 100) may cancel out,resulting in the following:

I _(REF)/(I _(REF) +I _(TH))−(1/R _(REF))/((1/R _(REF))+(1/R_(TH))).  (Eq. Eight)

Multiplying the numerator and the denominator of equation eight byR_(REF)*R_(TH) shows that the current ratio in equation eight may beequivalent to the following resistor ratio:

F=R _(TH)/(R _(TH) +R _(REF))  (Eq. Nine)

where Γ represents the resistance ratio, R_(REF) represents theresistance of reference resistor 106, and R_(TH) represents theresistance of thermistor 105.

Reference resistor 106 may be a discrete off-chip component that mayhave approximately the same resistance value across a temperature rangeof, for example, eighty-five to negative thirty degrees Celsius. On theother hand, thermistor 105 may have a resistance that may vary by designacross such a temperature range. Accordingly, the value of theresistance ratio may vary as a function of temperature across thetemperature range. Look-up map 380 may be configured to receive theresistance ratio from divider 370 and to output a temperature valuebased on the resistance ratio. In some embodiments, look-up map 380 mayinclude a non-volatile memory including table of potential resistanceratios and corresponding temperature values across a temperature range.For such embodiments, look-up map 380 may receive a resistance ratiofrom divider 370, determine the closest resistance-ratio entry in thetable, and output the temperature that corresponds to the closestresistance-ratio entry in the table. The resolution of the temperatureoutput for such embodiments may depend the number of potentialresistance-ratio values in such a table of resistance-ratio values. Forexample, look-up map 380 may include a table with one-hundred andsixteen entries in order to provide a resolution of one-degree Celsiusover a potential range of eighty-five degrees Celsius to negative thirtydegrees Celsius.

In some embodiments, look-up map 380 may be configured to interpolate atemperature value based on two or more table entries. For example, if aresistance-ratio input is half way between the resistance ratios of twotable entries, look-up map 380 may calculate a temperature that may behalf way between the corresponding temperature output values for the twotable entries. In some embodiments, look-up map 380 may include analgorithm instead of a table of resistance ratios and correspondingtemperature values. For such embodiments, look-up map 380 may calculatea temperature output based on the resistance ratio and the temperaturealgorithm. The resistance-ratio and temperature values stored in a tablein look-up map 380, and/or any parameters used in an algorithm oflook-up map 380, may be based on known characteristics for thermistor105 and/or reference resistor 106.

Temperature measurement system 300 may be configured to measure andoutput temperatures across any suitable temperature range for a givenapplication. For example, in consumer electronic applications,temperature measurement system 300 may be configured to measure andoutput temperature values from eighty-five to negative thirty degreesCelsius. As another example, in automotive applications, temperaturemeasurement system 300 may be configured to measure and outputtemperature values from one-hundred-and-forty to negative eighty-fivedegrees Celsius.

Because the final temperature measurement may be based on a ratioincluding R_(TH) and R_(REF), the accuracy of the final temperaturemeasurement may depend on the relative value of R_(TH) as compared toR_(REF), rather than the accuracy of R_(TH) or R_(REF) individually.Various design parameters for ADC 310 and decimators 340 (e.g., thenumber of cycles of sigma-delta operation, the gain of ADC 310, and theorder of decimation) may have the same impact on the respectiveaccuracies of R_(TH) and R_(REF). Thus, while such design parameters mayimpact the measurement of R_(TH) and/or R_(REF) individually, thosedesign parameters may have only a negligible impact on the resistanceratio including R_(TH) and R_(REF). As a result, temperature measurementsystem 300 may achieve a high degree of accuracy without tuning the gainof ADC 310 and/or performing a normalization on ADC 310 and decimators340.

For similar reasons, temperature measurement system 300 may avoid errorscaused by a gain error in buffer 100. For example, if semiconductordevice mismatch causes the gain of buffer 100 to be five-percent largerthan designed, the same five percent error may be incurred by each ofmeasurements M1 and M2 for reference resistor 106, and each ofmeasurements M3 and M4 for thermistor 105. In such situations, R_(TH)and R_(REF) may both include a five percent error. However, because thefive percent error may effect R_(TH) and R_(REF) equally, such an errormay cancel out of a resistance ratio including R_(TH) and R_(REF).Accordingly, the temperature measurement output may be unaffected by again error of buffer 100.

Because the architecture of temperature measurement system 300 may allowfor potential gain errors in buffer 100 and/or ADC 310 to be canceledout, buffer 100 and/or ADC 310 may be designed with relaxed requirementsfor various parameters that may contribute to such gain errors. Forexample, as described above with reference to FIG. 1, the gain of buffer100 may be affected by the size of PMOS 132 as compared to the size ofPMOS 130. The ratio of the sizes of PMOS 130 and PMOS 132 may beaffected, for example, by various semiconductor processing defects ormismatch. Transistors with smaller dimensions (e.g., channel width andchannel length for NMOS or PMOS devices) may be more susceptible tosemiconductor processing defects and/or mismatch than transistors withlarger dimensions. However, because any mismatch-induced gain error maybe cancelled out at a later stage, PMOS 130 and PMOS 132 may beconfigured with nominal dimensions. Thus, a significant amount ofsemiconductor area may be saved.

Though the resistance ratio may be described above asΓ=R_(TH)/(R_(TH)+R_(REF)), digital calculation stage 315 may beconfigured to implement any suitable ratio including R_(TH) and R_(REF).For example, DEMUX 330 may be configured to route measurements M1 and M2to decimators 340 c and 340 d respectively, and to route measurements M3and M4 to decimators 340 a and 340 b respectively. In such embodiments,adder 360 and divider 370 may combine to calculate a resistance ratio ofR_(REF)/(R_(REF)+R_(TH)) rather than R_(TH)/(R_(TH)+R_(REF)). In someembodiments, the functionality of adder 360 may be bypassed, and a ratioof R_(REF)/R_(TH) or R_(TH)/R_(REF) may be utilized.

FIG. 4 depicts a flow chart of an example method 400 for measuringtemperature, in accordance with the teachings of the present disclosure.At step 402, a first current signal based on a resistance of a resistormay be generated. For example, buffer 100 may generate a differentialoutput current that may be based on reference resistor 106. In someembodiments, buffer 100 may be set to a first polarity state during step402. At step 404, the first current signal may be converted into a firstdigital signal. For example, ADC 310 may be a sigma-delta ADC, and mayconvert the differential output current from buffer 100 into a stream ofdigital bits.

At step 406, a first complementary current signal based on theresistance of the resistor may be generated. For example, the polarityof buffer 100 may be changed from a first polarity state to a secondpolarity state, and buffer 100 may generate a differential outputcurrent that may be based on reference resistor 106. Accordingly, offsetcurrent (e.g., I_(OFF)) incurred during step 402 may be matched by anequivalent offset during step 406. At step 408, the first complementarycurrent signal may be converted into a first complementary digitalsignal. For example, ADC 310 may be a sigma-delta ADC, and may convertthe differential output current from buffer 100 into a stream of digitalbits.

At step 412, a second current signal based on a resistance of athermistor may be generated. For example, buffer 100 may generate adifferential output current that may be based on thermistor 105. In someembodiments, buffer 100 may be in a first polarity state during step412. At step 414, the second current signal may be converted into asecond digital signal. For example, ADC 310 may be a sigma-delta ADC,and may convert the differential output current from buffer 100 into astream of digital bits.

At step 416, a second complementary current signal based on theresistance of the thermistor may be generated. For example, the polarityof buffer 100 may be changed from a first polarity state to a secondpolarity state, and buffer 100 may generate a differential outputcurrent that may be based on thermistor 105. Accordingly, offset current(e.g., I_(OFF)) incurred during step 412 may be matched by an equivalentoffset during step 416. At step 418, the second complementary currentsignal may be converted into a second complementary digital signal. Forexample, ADC 310 may be a sigma-delta ADC, and may convert thedifferential output current from buffer 100 into a stream of digitalbits.

At step 420, a first digital value corresponding to the resistance ofthe resistor may be determined. In some embodiments, the first digitalvalue may be based on the first digital signal and the firstcomplementary digital signal. For example, DEMUX 330 may route the firstdigital signal (e.g., the stream of bits generated by ADC 310 duringstep 404) to decimator 340 a. Similarly, DEMUX 330 may route the firstcomplementary digital signal (e.g., the stream of bits generated by ADC310 during step 408) to decimator 340 b. Decimators 340 a and 340 b mayconvert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 a.Offset canceller 350 a may subtract the output of decimator 340 b fromthe output of decimator 340 a and output a digital value, I_(REF), whichmay depend on the resistance of reference resistor 106.

At step 422, a second digital value corresponding to the resistance ofthe thermistor may be determined. In some embodiments, the seconddigital value may be based on the second digital signal and the secondcomplementary digital signal. For example, DEMUX 330 may route thesecond digital signal (e.g., the stream of bits generated by ADC 310during step 414) to decimator 340 c. Similarly, DEMUX 330 may route thesecond complementary digital signal (e.g., the stream of bits generatedby ADC 310 during step 418) to decimator 340 d. Decimators 340 c and 340d may convert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 b.Offset canceller 350 b may subtract the output of decimator 340 d fromthe output of decimator 340 c and output a digital value, I_(TH), whichmay depend on the resistance of thermistor 105.

At step 424, a resistance ratio may be calculated based on the firstdigital value and the second digital value. For example, adder 360 anddivider 370 may combine to divide the first digital value (e.g.,I_(REF)) by the sum of the first digital value (e.g., I_(REF)) and thesecond digital value (e.g., I_(TH)). As shown in equations eight andnine above, a ratio such as I_(REF)/(I_(REF)+I_(TH)) may be equivalentto a resistance ratio such as R_(TH)/(R_(TH)+R_(REF)).

At step 426, a temperature output value may be determined based on theresistance ratio. For example, look-up map 380 may contain a table ofpotential resistance ratios and corresponding temperature output values.The resistance ratio from step 424 may used to look up the closestresistance-ratio entry in the table, and the corresponding temperatureoutput value may be returned.

Although FIG. 4 discloses a particular number of steps to be taken withrespect to method 400, method 400 may be executed with greater or lessersteps than those depicted in FIG. 4. For example, method 400 may beexecuted without steps 406, 408, 416, and 418. In addition, althoughFIG. 4 discloses a certain order of steps to be taken with respect tomethod 400, the steps included in method 400 may be completed in anysuitable order. For example, step 402 and step 404 may occursimultaneously.

FIG. 5 illustrates a block diagram depicting a calibrated temperaturemeasurement system 500, in accordance with the teachings of the presentdisclosure. Temperature measurement system 500 may include similarcomponents, and may perform similar measurements, as compared totemperature measurement system 300. Temperature measurement system 500may also include additional components that may provide for thecharacterization of an on-chip reference resistor (e.g., referenceresistor 506), and the calibration of temperature measurements that mayuse such an on-chip reference resistor.

In some embodiments, reference resistor 506 may be an on-chip resistorlocated on the same semiconductor chip as analog input stage 505 anddigital calculation stage 515. For example, reference resistor maycomprise a polysilicon structure located on a semiconductor chip, andmay be referred to as a “polysilicon resistor” or as a “poly resistor.”The value of poly resistors such as reference resistor 506 may vary froma designed resistance value based on semiconductor process variation.For example, the value of a poly resistor may vary up to plus or minusten percent due to semiconductor process variation. Such variation maybe more or less depending on the parameters of a given semiconductorprocess.

In order to account for the potential variation of reference resistor506, temperature measurement system 500 may be configured tocharacterize reference resistor 506 and then calibrate temperaturemeasurements that may be based in part on reference resistor 506. Forexample, prior to temperature measurement system 500 being implementedin a final product, the resistance of reference resistor 506 may becompared to the resistance of a highly accurate test resistor 507 in atest environment. In some embodiments, test resistor 507 may beconfigured to have a resistance equal to the desired resistance ofreference resistor 506. As described in further detail below, acharacterization of the resistance of reference resistor 506 as comparedto the resistance of test resistor 507 (e.g., the ideal resistance forreference resistor 506) may be stored in calibration module 585. Thecharacterization of reference resistor 506 may then be used to adjustany temperature measurements that may be performed based on referenceresistor 506 and thermistor 105 in a final product incorporatingtemperature measurement system 500.

Temperature measurement system 500 may include analog input stage 505.Like analog input stage 305, analog input stage 505 may include buffer100 and ADC 310. Because reference resistor 506 may be an on-chip devicewithin analog input stage 505, reference resistor 506 may be coupleddirectly to the internal multiplexors of buffer 100 without the use of apin. During temperature measurements, thermistor 105 may be coupled topin 502. But, during a characterization of reference resistor 506, testresistor 507 may be coupled to pin 502 instead of thermistor 105. Thecoupling of test resistor 507 to pin 502 may occur, for example, in atest environment prior to temperature measurement system 500 beingincorporated in a product with thermistor 105.

With test resistor 507 coupled to pin 502, analog input stage 505 mayperform a series of measurements in a similar manner as described abovefor analog input stage 305. Moreover, DEMUX 330, decimators 340 a-d, andoffset cancellers 350 a-b may convert those measurements into multi-bitvalues, I_(REF) and I_(TEST), in a similar manner as described above fordigital calculation stage 315.

For example, each of the four decimators 340 a-d may be configured toreceive the stream of digital bits from ADC 310 during one of fourmeasurements (e.g., measurements M1 through M4). DEMUX 330 may routemeasurement M1 to decimator 340 a, measurement M2 to decimator 340 b,measurement M3 to decimator 340 c, and measurement M4 to decimator 340d. Measurements M1 and M2 may include opposing measurements of referenceresistor 506. Measurement M1 may include the stream of bits from ADC 310during a first period of time when reference resistor 506 may bemeasured with buffer 100 set to a first polarity setting. In addition,measurement M2 may include the stream of bits from ADC 310 during asecond period of time when reference resistor 506 may be measured withbuffer 100 set to a second polarity setting. Similarly, measurements M3and M4 may include opposing measurements for test resistor 507. Forexample, measurement M3 may include the stream of bits from ADC 310during a third period of time when test resistor 507 may be measuredwith buffer 100 set to the first polarity setting. In addition,measurement M4 may include the stream of bits from ADC 310 during afourth period of time when test resistor 507 may be measured with buffer100 set to the second polarity setting.

Offset canceller 350 a may then combine the outputs of decimators 340 aand 340 b, and may output a multi-bit value, I_(REF), that maycorrespond to the resistance of reference resistor 506. Likewise, offsetcanceller 350 b may combine the outputs for decimators 340 c and 340 d,and may output a multi-bit value, I_(TEST), that may correspond to theresistance of test resistor 507. During characterization, thefunctionality of adder 360 may be bypassed. For example, I_(TEST) may berouted directly to divider 570 via path 565. I_(REF) may also be routedto divider 570. Because I_(TEST) and I_(REF) may be inverselyproportional to the respective resistances of test resistor 507 andreference resistor 506, a current ratio such as I_(TEST)/I_(REF) may beequivalent to a resistance ratio such as R_(REF)/R_(TEST).

In some embodiments, test resistor 507 may have a highly accurateresistance that may be approximately equal to the designed idealresistance of reference resistor 506. Accordingly, divider 570 maycalculate a characterization ratio between the actual resistance ofreference resistor 506 (e.g., R_(REF)) and the ideal resistance forreference resistor 506 (e.g., R_(TEST)). Such a characterization ratiomay be expressed as γ=R_(REF)/R_(TEST). The resistor characterizationinformation may then be stored in a memory. For example, the ratio ofR_(REF) divided by R_(TEST) may be stored in calibration module 585. Insome embodiments, the data used to calculate the resistance ratio may bestored in calibration module 585 in addition to or in place of thecharacterization ratio. In such embodiments, the stored characterizationinformation may used at a later time (e.g., during a temperaturemeasurement) to calculate the characterization ratio. Calibration module585 may include any type of non-volatile memory. For example,calibration module 585 may include a plurality of digital fuses that maybe physically burned in. In some embodiments, calibration module 585 mayinclude a plurality of EEPROM bits which may be electronicallyprogrammed. Accordingly, the ratio of R_(REF) divided by R_(TEST) may beprovided to look-up map 580 during subsequent temperature measurements.

After the performance of a characterization, test resistor 507 may bede-coupled from pin 502, and thermistor 105 may be coupled to pin 502.Temperature measurement system 500 may then perform measurements ofreference resistor 506 and thermistor 105 in a similar manner asdescribed above with reference to FIG. 3 for thermistor 105 andreference resistor 106. For example, analog input stage 505 may performtwo measurements with opposing polarities of buffer 100 for referenceresistor 506, and two measurements with opposing polarities of buffer100 for thermistor 105. DEMUX 330, decimators 340 a-d, and offsetcancellers 350 a-b may then convert those measurements into a multi-bitdigital value corresponding to the resistance of thermistor 105 (e.g.,I_(TH)), and a multi-bit digital value corresponding to the resistanceof reference resistor 506 (e.g., I_(REF)). Similar to the descriptionabove with reference to equations eight and nine, adder 360 and divider370 may then combine to calculate a resistance ratio that may berepresented by the formula:

Γ_(actual) =R _(REF)/(R _(REF) +R _(TH))  (Eq. Ten)

where Γ_(actual) may be the calculated actual resistance ratio, R_(REF)may represent the actual resistance of reference resistor 506, andR_(TH) may represent the resistance of thermistor 105. The actualresistance ratio, Γ_(actual), may then be provided to look-up map 580.

FIG. 6 illustrates a block diagram of look-up map 580, in accordancewith the teachings of the present disclosure. Similar to look-up map380, look-up map 580 may include a table 582 which may includeresistance-ratio entries and corresponding temperature values that maybe based on known characteristics of thermistor 105. Theresistance-ratio entries and the temperature values in table 582 mayalso be based on the ideal designed resistance of reference resistor506, which may not account for any process variation that may havevaried the actual resistance of reference resistor 506. Accordingly, theresistance-ratio entries in table 582 may be referred to as the idealresistance ratios (Γ_(ideal)). Look-up map 580 may also includeresistance ratio converter 584 that may be configured to calculateactual resistance ratios (Γ_(actual)) based on: (i) the ideal resistanceratio (Γ_(ideal)); and (ii) the resistor-characterization ratio (γ).

As shown by the following series of equations, Γ_(actual) may bedetermined as a function of Γ_(ideal) and γ.

Γ_(actual) =R _(REF)(R _(REF) +R _(TH));  (Eq. Eleven)

Equation eleven may be re-written as:

Γ_(actual)=1/(1+(R _(TH) /R _(REF)));  (Eq. Twelve)

Solving for the ratio of the actual resistances, the following may beobtained:

R _(TH) /R _(REF)=(1/Γ_(actual))−1;  (Eq. Thirteen)

Because the ideal resistance for R_(REF) may be the resistance ofR_(TEST), the ideal resistance ratio may be expressed in a similarmanner as equation thirteen as follows:

R _(TH) /R _(TEST)=(1/Γ_(ideal))−1;  (Eq. Fourteen)

Multiplying both sides of equation fourteen by (1/γ) may result in:

R _(TH)(γ*R _(TEST))=((1/Γ_(ideal))−1)/γ;  (Eq. Fifteen)

Substituting R_(REF)=γ*R_(TEST) into equation fourteen may result in:

R _(TH) /R _(REF)=((1/Γ_(ideal))−1)/γ;  (Eq. Sixteen)

And substituting equation sixteen into equation twelve, the followingequation for Γ_(actual) as a function of Γ_(ideal) and γ may beobtained:

Γ_(actual)=1/(1+(((1/Γ_(deal))−1)/γ)).  (Eq. Seventeen)

In some embodiments, resistance ratio converter 584 may calculate avalue of Γ_(actual) for every entry in table 582. Accordingly,calibrated table 586 may include an Γ_(actual) value and a correspondingtemperature value for each temperature value that may be contained intable 582. When look-up map 580 receives a resistance ratio from divider570, look-up map 580 may determine the closest Γ_(actual) entry incalibrated table 586, and may return the corresponding temperatureoutput value. The resolution of the temperature measurement may dependon the number of Γ_(actual) values in calibrated table 586. For example,calibrated table 586 may include one-hundred and sixteen entries inorder to provide a resolution of one-degree Celsius over a range ofeighty-five degrees Celsius to negative thirty degrees Celsius.

In some embodiments, look-up map 580 may be configured to interpolate atemperature value based on two or more entries in calibrated table 586.For example, if a resistance-ratio input is half way between twoΓ_(actual) values in table 586, look-up map 580 may calculate and outputa temperature that may be half way between the corresponding temperaturevalues for the two table entries. In some embodiments, look-up map 580may include an algorithm instead of a table of resistance ratios andcorresponding temperature values. For such embodiments, look-up map 380may calculate a temperature output based on the actual resistance ratio,the resistor-characterization ratio, and known characteristics ofthermistor 105.

The characterization of on-chip reference resistor 506 and thecalibration of temperature measurements that may be based on referenceresistor 506 may allow for any reference-resistor errors to be minimizedand for costs to be reduced. For example, the monetary cost of anexternal reference resistor with a one-percent accuracy rating may besignificantly more than the monetary costs of the incrementalsemiconductor space used to incorporate reference resistor 506 on thesame chip as analog input stage 505 and/or digital calculation stage515. Moreover, calibrating temperature measurements with thecharacterization information for reference resistor 506 may achieve ahigher degree of accuracy (e.g., 0.1%) than possible with, for example,a one-percent off-chip reference resistor. Additionally, the on-chipincorporation of reference resistor 506 may reduce the number of pinsrequired for a given application incorporating a temperature measurementsystem. Accordingly, the semiconductor packaging costs may be reducedand the complexity of a printed-circuit board layout for an applicationincluding temperature measurement system 500 may be simplified.

FIG. 7 depicts a flow chart of an example method 700 for calibratingtemperature measurement system 500, in accordance with the teachings ofthe present disclosure.

At step 702, a first current signal based on a resistance of a referenceresistor may be generated. For example, buffer 100 may generate adifferential output current that may be based on reference resistor 506.In some embodiments, buffer 100 may be in a first polarity state duringstep 702. At step 704, the first current signal may be converted into afirst digital signal. For example, ADC 310 may be a sigma-delta ADC, andmay convert the differential output current from buffer 100 into astream of digital bits.

At step 706, a first complementary current signal based on theresistance of the reference resistor may be generated. For example, thepolarity of buffer 100 may be changed from a first polarity state to asecond polarity state, and buffer 100 may generate a differential outputcurrent that may be based on reference resistor 506. Accordingly, offsetcurrent (e.g., I_(OFF)) incurred during step 702 may be matched by anequivalent offset during step 706. At step 708, the first complementarycurrent signal may be converted into a first complementary digitalsignal. For example, ADC 310 may be a sigma-delta ADC, and may convertthe differential output current from buffer 100 into a stream of digitalbits.

At step 712, a second current signal based on a resistance of a testresistor may be generated. For example, and buffer 100 may generate adifferential output current that may be based on test resistor 507. Insome embodiments, buffer 100 may be in a first polarity state duringstep 712. At step 714, the second current signal may be converted into asecond digital signal. For example, ADC 310 may be a sigma-delta ADC,and may convert the differential output current from buffer 100 into astream of digital bits.

At step 716, a second complementary current signal based on theresistance of the test resistor may be generated. For example, thepolarity of buffer 100 may be changed from a first polarity state to asecond polarity state, and buffer 100 may generate a differential outputcurrent that may be based on test resistor 507. Accordingly, offsetcurrent (e.g., I_(OFF)) incurred during step 712 may be matched by anequivalent offset during step 716. At step 718, the second complementarycurrent signal may be converted into a second complementary digitalsignal. For example, ADC 310 may be a sigma-delta ADC, and may convertthe differential output current from buffer 100 into a stream of digitalbits.

At step 720, a first digital value corresponding to the resistance ofthe reference resistor may be determined. In some embodiments, the firstdigital value may be based on the first digital signal and the firstcomplementary digital signal. For example, DEMUX 330 may route the firstdigital signal (e.g., the stream of bits generated by ADC 310 duringstep 704) to decimator 340 a. Similarly, DEMUX 330 may route the firstcomplementary digital signal (e.g., the stream of bits generated by ADC310 during step 708) to decimator 340 b. Decimators 340 a and 340 b mayconvert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 a.Offset canceller 350 a may subtract the output of decimator 340 b fromthe output of decimator 340 a and may output a digital value (e.g.,I_(REF)) that may correspond to the resistance of reference resistor506.

At step 722, a second digital value corresponding to the resistance ofthe test resistor may be determined. In some embodiments, the seconddigital value may be based on the second digital signal and the secondcomplementary digital signal. For example, DEMUX 330 may route thesecond digital signal (e.g., the stream of bits generated by ADC 310during step 714) to decimator 340 c. Similarly, DEMUX 330 may route thesecond complementary digital signal (e.g., the stream of bits generatedby ADC 310 during step 718) to decimator 340 d. Decimators 340 c and 340d may convert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 b.Offset canceller 350 b may subtract the output of decimator 340 d fromthe output of decimator 340 c and output a digital value and may outputa digital value (e.g., I_(TEST)) that may correspond to the resistanceof test resistor 507.

At step 724, a resistor-characterization ratio may be calculated basedon the first digital value and the second digital value. For example,divider 370 may divide the second digital value (e.g., I_(TEST)) by thefirst digital value (e.g., I_(REF)) to get a value that may beequivalent to a ratio of the resistance of reference resistor 506divided test resistor 507 (e.g., R_(REF)/R_(TEST)). Similar to thedescription above with reference to equation seven, dividing one digitalvalue (e.g., I_(TEST)) by another digital value (e.g., I_(REF)) maycancel the voltage offset of amplifier 120 in buffer 100. At step 726,the resistor-characterization ratio may be stored in a memory. In someembodiments, the memory may be a non-volatile memory, and the storedresistor-characterization ratio may be available to temperaturemeasurement system 500 during later performed temperature measurements.

Although FIG. 7 discloses a particular number of steps to be taken withrespect to method 700, method 700 may be executed with greater or lessersteps than those depicted in FIG. 7. For example, method 700 may beexecuted without steps 706, 708, 716, and 718. In addition, althoughFIG. 7 discloses a certain order of steps to be taken with respect tomethod 700, the steps included in method 700 may be completed in anysuitable order. For example, step 702 and step 704 may occursimultaneously.

FIG. 8 depicts a flow chart of an example method 800 for measuringtemperature, in accordance with the teachings of the present disclosure.

At step 802, a first current signal based on a resistance of a resistormay be generated. For example, buffer 100 may generate a differentialoutput current that may be based on reference resistor 506. In someembodiments, buffer 100 may be in a first polarity state during step802. At step 804, the first current signal may be converted into a firstdigital signal. For example, ADC 310 may be a sigma-delta ADC, and mayconvert the differential output current from buffer 100 into a stream ofdigital bits.

At step 806, a first complementary current signal based on theresistance of the resistor may be generated. For example, the polarityof buffer 100 may be changed from a first polarity state to a secondpolarity state, and buffer 100 may generate a differential outputcurrent that may be based on reference resistor 506. Accordingly,current offset (e.g., I_(OFF)) incurred during step 802 may be matchedby an equivalent offset during step 806. At step 808, the firstcomplementary current signal may be converted into a first complementarydigital signal. For example, ADC 310 may be a sigma-delta ADC, and mayconvert the differential output current from buffer 100 into a stream ofdigital bits

At step 812, a second current signal based on a resistance of athermistor may be generated. For example, buffer 100 may generate adifferential output current that may be based on thermistor 105. In someembodiments, buffer 100 may be in a first polarity state during step812. At step 814, the second current signal may be converted into asecond digital signal. For example, ADC 310 may be a sigma-delta ADC,and may convert the differential output current from buffer 100 into astream of digital bits.

At step 816, a second complementary current signal based on theresistance of the thermistor may be generated. For example, the polarityof buffer 100 may be changed from a first polarity state to a secondpolarity state, and buffer 100 may generate a differential outputcurrent that may be based on thermistor 105. Accordingly, offset current(e.g., I_(OFF)) incurred during step 812 may be matched by andequivalent offset during step 816. At step 818, the second complementarycurrent signal may be converted into a second complementary digitalsignal. For example, ADC 310 may be a sigma-delta ADC, and may convertthe differential output current from buffer 100 into a stream of digitalbits.

At step 820, a first digital value corresponding to the resistance ofthe reference resistor may be determined. In some embodiments, the firstdigital value may be based on the first digital signal and the firstcomplementary digital signal. For example, DEMUX 330 may route the firstdigital signal (e.g., the stream of bits generated by ADC 310 duringstep 804) to decimator 340 a. Similarly, DEMUX 330 may route the firstcomplementary digital signal (e.g., the stream of bits generated by ADC310 during step 808) to decimator 340 b. Decimators 340 a and 340 b mayconvert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 a.Offset canceller 350 a may subtract the output of decimator 340 b fromthe output of decimator 340 a and output a digital value (e.g., I_(REF))that may correspond to the resistance of reference resistor 506.

At step 822, a second digital value corresponding to the resistance ofthe thermistor may be determined. In some embodiments, the seconddigital value may be based on the second digital signal and the secondcomplementary digital signal. For example, DEMUX 330 may route thesecond digital signal (e.g., the stream of bits generated by ADC 310during step 814) to decimator 340 c. Similarly, DEMUX 330 may route thesecond complementary digital signal (e.g., the stream of bits generatedby ADC 310 during step 818) to decimator 340 d. Decimators 340 c and 340d may convert their respectively received digital signals into multi-bitvalues, and may send those multi-bit values to offset canceller 350 b.Offset canceller 350 b may subtract the output of decimator 340 d fromthe output of decimator 340 c and output a digital value (e.g., I_(TH))that corresponds to the resistance of thermistor 105.

At step 824, a resistance ratio may be calculated based on the firstdigital value and the second digital value. For example, adder 360 anddivider 370 may combine to divide the first digital value (e.g.,I_(REF)) by the sum of the first digital value (e.g., I_(REF)) and thesecond digital value (e.g., I_(TH)). As described above with referenceto equation seven, dividing the first digital value (e.g., I_(REF)) bythe sum of the first digital value (e.g., I_(REF)) and the seconddigital value (e.g., I_(TH)) may cancel the voltage offset of amplifier120 in buffer 100. Further, as shown above with reference to equationseight and nine, such a ratio of current values may be equivalent to aresistance ratio including the resistance of reference resistor 506 andthermistor 105. At step 826, a temperature output value may bedetermined based on the resistance ratio and theresistor-characterization ratio.

Although FIG. 8 discloses a particular number of steps to be taken withrespect to method 800, method 800 may be executed with greater or lessersteps than those depicted in FIG. 8. For example, method 800 may beexecuted without steps 806, 808, 816, and 818. In addition, althoughFIG. 8 discloses a certain order of steps to be taken with respect tomethod 800, the steps included in method 800 may be completed in anysuitable order. For example, step 802 and step 804 may occursimultaneously.

FIG. 9 illustrates a schematic diagram of a current-mode buffer 900, inaccordance with the teachings of the present disclosure. Buffer 900 maybe an input stage for a system that, as described in greater detailbelow with reference to FIG. 11, may be configured to measure voltage.

Buffer 900 may include a common-mode voltage reference (V_(CM)), anamplifier 920, PMOS transistors 130, 132, 134, NMOS transistors 140,142, 144, as well as switch 112 and switch 912. Buffer 100 may beconfigured to sense a voltage, and to output a current that may be basedon the sensed voltage. In some embodiments, the output current may beproportional to the sensed voltage.

In some embodiments, amplifier 920 may have a negative input coupled toone out of a plurality of voltage inputs. For example, the negativeinput of amplifier 920 may be coupled to one out of V_(CM) and V_(X) viaswitch 912. In a first state, switch 912 may couple V_(CM) to amplifier920, and in a second state, switch 912 may coupled V_(X) to amplifier920. The output of amplifier 920 may be coupled to feedback node 125,which may drive the gate of PMOS 130. PMOS 130 may in turn provide afeedback current (I_(FB)) to resistor 906, which may be coupled to apositive input of amplifier 920. In some embodiments, resistor 906 maybe an off-chip device, and in some embodiments, resistor 906 may be onon-chip device located on the same semiconductor chip as buffer 900. Thefeedback loop formed by PMOS 130 may drive a feedback current sufficientto force the voltage at the positive input to be equivalent to thevoltage coupled to the negative input of amplifier 920. Accordingly,when the V_(CM) is coupled to the negative input of amplifier 920, thefeedback current may be expressed as I_(FB)=V_(CM)/R₉₀₆, where R₉₀₆ maybe the resistance of resistor 906. Likewise, when the V_(X) is coupledto the negative input of amplifier 920, the feedback current may beexpressed as I_(FB)=V_(X)/R₉₀₆.

In some embodiments, buffer 900 may be configured to generate an outputcurrent in a similar manner as described above with reference to FIG. 1Afor buffer 100. For example, PMOS 132 may be configured to mirror PMOS130 at any suitable ratio, and thus may generate a single-ended outputcurrent, I_(OUT), that may be proportional to I_(FB). In someembodiments, PMOS 130 may be coupled to switch 112, which may routeI_(OUT) to one of two output nodes. For example, when switch 112 is in afirst state, switch 112 may route I_(OUT) to positive output node 954.When switch 112 is in a second state, switch 112 may route I_(OUT) tonegative output node 952.

In some embodiments, buffer 900 may be configured to convert thesingle-ended output current I_(OUT) into a differential output currentby sinking a current of one-half I_(OUT) from both the positive outputnode 954 and the negative output node 952. For example, PMOS 134 may beconfigured to mirror PMOS 130, but at one-half the ratio with which PMOS132 may mirror PMOS 130. Accordingly, PMOS 132 may generate asingle-ended current equivalent to one-half I_(OUT). NMOS 140 may beconfigured to accept the one-half I_(OUT) current. NMOS 140 may beself-biased (i.e., have a gate coupled to its drain) and may generate agate bias for NMOS 142 and NMOS 144. NMOS 142 and NMOS 144 may have agate coupled to the gate of NMOS 140 and may be configured to have thesame size as NMOS 140. Accordingly, NMOS 142 and NMOS 144 may each beconfigured to mirror the one-half I_(OUT) current of NMOS 140. In someembodiments, NMOS 142 may have a drain coupled to negative output node952 and NMOS 144 may have a drain coupled to positive output node 154.Accordingly, NMOS 142 may sink a current of approximately one-halfI_(OUT) from negative output node 952, and NMOS 144 may sink a currentof approximately one-half I_(OUT) from positive output node 954.

In addition to alternating the output polarity of buffer 900 byalternating the state of switch 112, buffer 900 may be configured toalternate the input polarity by alternating the input polarity state ofamplifier 920. The alternating polarity states of amplifier 920 aredescribed in further detail below with reference to FIG. 10.

FIG. 10 illustrates a schematic diagram of amplifier 920, in accordancewith the teachings of the present disclosure. Amplifier 920 may includea positive input (V_(IN) ⁺), a negative input (V_(IN) ⁻), currentsources 931 and 932, switches 913 a-d, a differential pair formed byPMOS 933 and PMOS 934, a pair of matched transistors formed by NMOS 935and NMOS 936, and a second-stage formed by NMOS 937 and miller capacitor938.

In some embodiments, amplifier 920 may be configured to alternatepolarity states based on the state of switches 913 a-d. For example,PMOS 933 and PMOS 934 may be configured as an input-stage differentialpair biased by current source 931. In a first polarity state ofamplifier 920, V_(IN) ⁺ may be coupled to the gate of PMOS 933 by switch913 a, and V_(IN) ⁻ may be coupled to the gate of PMOS 934 by switch 913b. The drain of PMOS 933 may be coupled to the drain of NMOS 935.Likewise, the drain of PMOS 934 may be coupled to the drain of NMOS 936.In the first polarity state, switch 913 d may couple the drain of NMOS936 to the gate of NMOS 936 and the gate of NMOS 935, making NMOS 936 aself-biased device that also biases NMOS 935. Accordingly, the output ofthe first stage may be the node coupling the drain of PMOS 933 and thedrain of NMOS 935. In turn, this first stage output node may be coupledby switch 913 c to the gate NMOS 937 in the second stage of amplifier920. The drain of NMOS 937 may be coupled to current source 932 at theoutput node (OUT) of amplifier 920.

In a second polarity state of amplifier 920, the state of each of theswitches 913 a-d may be alternated to a second state. For example,V_(IN) ⁻ may be coupled to the gate of PMOS 933 by switch 913 a, andV_(IN) ⁺ may be coupled to the gate of PMOS 934 by switch 913 b. Thedrain of PMOS 933 may be coupled to the drain of NMOS 935. Likewise, thedrain of PMOS 934 may be coupled to the drain of NMOS 936. In the secondpolarity state, switch 913 d may couple the drain of NMOS 935 to thegate of NMOS 935 and the gate of NMOS 936, making NMOS 935 a self-biaseddevice that also biases NMOS 936. Accordingly, the output of the firststage may be the node coupling the drain of PMOS 934 and the drain ofNMOS 936. In turn, this first stage output node may be coupled by switch913 c to NMOS 937 of the second stage of amplifier 920. The drain ofNMOS 937 may be coupled to current source 932 at the output node (OUT)of amplifier 920. Miller capacitor 938 may be coupled from the gate ofNMOS 937 to the drain of NMOS 937, and may have a value that maydetermine the unity gain frequency and the phase margin of amplifier 920in both the first polarity state and the second polarity state.

Referring back to FIG. 9, buffer 900 may operate in one of fourpotential states at a time. For example, output switch 112 may operatein either a first or second output-switch state. For the purposes of thepresent disclosure, the alternating of output switch 112 may be referredto herein as alternating the output polarity of buffer 900. In addition,amplifier 920 may operate in either a first or second polarity state.For the purposes of the present disclosure, alternating the polaritystate of amplifier 920 may be referred to herein as alternating theinput polarity of buffer 900. With two input polarity states and twooutput polarity states, buffer 900 may operate in one of four potentialstates. Moreover, output switch 912 may couple either V_(CM) or V_(X) tothe negative input of amplifier 920. Accordingly, buffer 900 may operatein one of four states at a time while measuring V_(CM), and one of fourstates at a time while measuring V_(X), for a combined total of eightunique current-mode measurements. The use of those eight measurements to(i) cancel any offsets incurred in buffer 900 or incurred downstreamfrom buffer 900; and (ii) calculate a value of V_(X) based on a knownvalue for V_(CM), is described in further detail below with reference toFIG. 11.

FIG. 11 illustrates a block diagram depicting voltage-measurement system950, in accordance with the teachings of the present disclosure.Voltage-measurement system 950 may include analog input stage 955 anddigital calculation stage 956. In some embodiments, analog input stage955 and digital calculation stage 956 may be included on a singlesemiconductor chip. In some embodiments, resistor 906 of analog inputstage 955 may be included on the same semiconductor chip as otherportions of analog input stage 955, and in some embodiments, resistor906 may be an external component.

Analog input stage 955 may include buffer 900 and ADC 310. As describedabove buffer 900 may be configured to output a differential currentsignal that may be proportional a selected one of V_(X) and V_(CM).Moreover, as described above, buffer 900 may alternate between two inputpolarity states and two output polarity states. For example, buffer 900may perform four measurements of V_(X) at time periods one through four,and four measurements of V_(CM) at time periods five through eight. Inturn, ADC 310 may convert each of the eight differential current signalsfrom buffer 900 into eight respective streams of digital bits. Theoutput of ADC 310 may in turn be communicated to digital calculationstage 956.

Digital calculation stage 956 may include digital logic configured toreceive and process one or more streams of digital bits from ADC 310 andto calculate a voltage and/or a voltage ratio. Digital calculation stage956 may include DEMUX 958, decimators 340 a-h, subtractors 960 a-d,adders 961 a-b, and divider 970. Digital calculation stage 956 mayinclude logic implemented in any suitable manner. For example, the logicof digital calculation stage 956 may be implemented in anapplication-specific integrated circuit (ASIC), in a field-programmablegate array (FPGA), in program instructions stored in a memory andconfigured to be executed by a multi-purpose processor, or any suitablecombination thereof.

In some embodiments, the input of digital calculation stage 956 may becoupled to the input of DEMUX 958. DEMUX 958 may be an eight-to-onedemultiplexor and may be configured to couple the input of digitalcalculation stage 956 to one of eight decimators 340 at a time. Eightdifferent measurements for V_(X) and V_(CM) performed by buffer 900,converted into digital form by ADC 310, and routed to decimators 340a-h, may be represented as described below with reference to equationseighteen through twenty-five.

A first measurement M1 may be routed to decimator 340 a. M1 may be basedon a current-mode measurement (and subsequent analog-to-digitalconversion) of V_(X) with buffer 900 in a first input polarity state anda first output polarity state. The first measurement M1, may berepresented as:

M1=((V _(X) +V _(OFF))/R ₉₀₆)+I _(OFF)  (Eq. Eighteen)

where V_(OFF) represents the input offset voltage of amplifier 920,I_(OFF) represents the input offset current of ADC 310, and R₉₀₆represents the resistance of resistor 906.

A second measurement M2 may be routed to decimator 340 b. M2 may bebased on a current-mode measurement (and subsequent analog-to-digitalconversion) of V_(X) with buffer 900 in a first input polarity state anda second output polarity state. The second measurement M2, may berepresented as:

M2=(−(V _(X) +V _(OFF))/R ₉₀₆)+I _(OFF).  (Eq. Nineteen)

As shown in equation nineteen, the current value represented by−(V_(X)+V_(OFF))/R₉₀₆ may be inverted as compared to equation eighteenbecause, in a second output polarity state, buffer 900 may alternate thepolarity of the current that may be routed to the positive and negativeoutput terminals of buffer 900, and thereby may alternate the polarityof the current that may output to ADC 310.

A third measurement M3 may be routed to decimator 340 c. M3 may be basedon a current-mode measurement (and subsequent analog-to-digitalconversion) of V_(X) with buffer 900 in a second input polarity stateand a first output polarity state. The third measurement M3, may berepresented as:

M3=((V _(X) −V _(OFF))/R ₉₀₆)+I _(OFF).  (Eq. Twenty)

As shown in equation twenty, V_(OFF) may be subtracted from V_(X)instead of added to V_(X) because any offset incurred during a firstinput polarity state (e.g., during measurement M1) may be invertedduring a second input polarity state.

A fourth measurement M4 may be routed to decimator 340 d. M4 may bebased on a current-mode measurement (and subsequent analog-to-digitalconversion) of V_(X) with buffer 900 in a second input polarity stateand a second output polarity state. The fourth measurement M4, may berepresented as:

M4=(−(V _(X) −V _(OFF))/R ₉₀₆)+I _(OFF).  (Eq. Twenty-One)

Measurements five through eight, M5-M8, may be performed on V_(CM) in asimilar manner as M1-M4 were performed on V_(X), and may be routed todecimators 340 e-h respectively. Measurements five through eight, may berepresented as follows:

M5=((V _(CM) +V _(OFF))/R ₉₀₆)+I _(OFF);  (Eq. Twenty-Two)

M6=(−(V _(CM) +V _(OFF))/R ₉₀₆)+I _(OFF);  (Eq. Twenty-Three)

M7=((V _(CM) −V _(OFF))/R ₉₀₆)+I _(OFF);  (Eq. Twenty-Four)

M8=(−(V _(CM) −V _(OFF))/R ₉₀₆)+I _(OFF).  (Eq. Twenty-Five)

As described above with reference to FIG. 3, each decimator 340 mayconvert a stream of digital bits received from ADC 310 into a multi-bitdigital value. Subtractors 960 a-d, adders 961 a-b, and divider 970 mayin turn further process the multi-bit digital values output bydecimators 340 a-h. For example, subtractor 960 a may subtract theoutput of decimator 340 b (e.g., a converted M2) from the output ofdecimator 340 a (e.g., a converted M1). Combining equations eighteen andnineteen, the output of subtractor 960 a may be represented as:

M1−M2=2*(V _(X) +V _(OFF))/R ₉₀₆.  (Eq. Twenty-Six)

As shown in equation twenty-six, I_(OFF) (e.g., the input offset currentof ADC 310) in equation eighteen and nineteen may cancel out.

Subtractor 960 b may subtract the output of decimator 340 d (e.g., aconverted M4) from the output of decimator 340 c (e.g., a converted M3).Combining equations twenty and twenty-one, the output of subtractor 960b may be represented as:

M3−M4=2*(V _(X) −V _(OFF))/R ₉₀₆.  (Eq. Twenty-Seven)

Subtractor 960 c may subtract the output of decimator 340 f (e.g., aconverted M6) from the output of decimator 340 e (e.g., a converted M5).Combining equations twenty-two and twenty-three, the output ofsubtractor 960 c may be represented as:

M5−M6=2*(V _(CM) +V _(OFF))/R ₉₀₆.  (Eq. Twenty-Eight)

Subtractor 960 d may subtract the output of decimator 340 h (e.g., aconverted M8) from the output of decimator 340 g (e.g., a converted M7).Combining equations twenty-four and twenty-five, the output ofsubtractor 960 d may be represented as:

M7−M8=2*(V _(CM) −V _(OFF))/R ₉₀₆.  (Eq. Twenty-Nine)

Adders 961 a and 961 b may in turn combine the outputs of subtractors960 a-d. For example, adder 961 a may add the output of subtractor 960 bto the output of subtractor 960 a. Combining equations twenty-six andtwenty-seven, the output of adder 961 a may be represented as:

(M1−M2)+(M3−M4)=(4*V _(X))/R ₉₀₆.  (Eq. Thirty)

As shown in equation twenty-nine, V_(OFF) (e.g., the input offsetvoltage of amplifier 920) may be cancelled out when combining theoutputs of subtractor 960 a and subtractor 960 b.

Adder 961 b may add the output of subtractor 960 d to the output ofsubtractor 960 c. Combining equations twenty-eight and twenty-nine, theoutput of adder 961 b may be represented as:

(M5−M6)+(M7−M8)=(4*V _(CM))/R ₉₀₆.  (Eq. Thirty-One)

As shown in equation thirty-one, V_(OFF) (e.g., the input offset voltageof amplifier 920) may be cancelled for the measurements of V_(CM) in asimilar manner as shown in equation thirty for the measurements ofV_(X).

The respective outputs of adders 961 a and 961 b may be communicated tothe inputs of divider 970. Divider 970 may divide one value by theother. For example, divider 970 may divide the output of adder 961 a bythe output of adder 961 b. Combining equations thirty and thirty-one,the output of divider 970 may be represented as:

((4*V _(X))/R ₉₀₆)/((4*V _(CM))/R ₉₀₆)=V _(X) /V _(CM).  (Eq.Thirty-Two)

As shown in equation thirty-two, the multiple of four and the value ofresistor 906 may cancel out, and the output of divider 970 may beequivalent to the ratio of V_(X) divided by V_(CM). Accordingly, theoutput of divider 970 may be referred to as a voltage ratio.

In some embodiments, the voltage of V_(CM) may be a known value. Forexample, V_(CM) may be a known value based on a bandgap voltage.Accordingly, the value for V_(X) may be determined based on the knownvalue of V_(CM) and the calculated voltage ratio V_(X)/V_(CM). Such adetermination may be performed in any suitable manner. In someembodiments, the output of divider 970 may be provided to a look-up mapwhich may include a table of voltage-ratio entries and correspondingoutput values for V_(X). In some embodiments, the value of V_(X) may bedetermined based on an algorithm rather than a look-up table. Forexample, the calculated ratio V_(X)/V_(CM) may be multiplied by a knownvalue of V_(CM) in order to obtain V_(X).

This disclosure encompasses all changes, substitutions, variations,alterations, and modifications to the example embodiments herein that aperson having ordinary skill in the art would comprehend. Similarly,where appropriate, the appended claims encompass all changes,substitutions, variations, alterations, and modifications to the exampleembodiments herein that a person having ordinary skill in the art wouldcomprehend. Moreover, reference in the appended claims to an apparatusor system or a component of an apparatus or system being adapted to,arranged to, capable of, configured to, enabled to, operable to, oroperative to perform a particular function encompasses that apparatus,system, component, whether or not it or that particular function isactivated, turned on, or unlocked, as long as that apparatus, system, orcomponent is so adapted, arranged, capable, configured, enabled,operable, or operative.

All examples and conditional language recited herein are intended forpedagogical purposes to aid the reader in understanding the principlesof the invention and the concepts contributed by the inventor tofurthering the art, and are to be construed as being without limitationto such specifically recited examples and conditions, nor does theorganization of such examples in the specification relate to a showingof the superiority and inferiority of the invention. Although theembodiments of the present inventions has been described in detail, itshould be understood that the various changes, substitutions, andalterations could be made hereto without departing from the spirit andscope of the invention.

What is claimed is:
 1. A temperature measurement system, comprising: aresistor; a thermistor; a resistance-to-current converter configured togenerate a current signal based on a resistance; an analog-to-digitalconverter (ADC) configured to: receive a first current signal based onthe resistor; convert the first current signal into a first digitalsignal; receive a second current signal based on the thermistor; andconvert the second current signal into a second digital signal; a memorycomprising resistor-characterization information; and a calculationstage communicatively coupled to an ADC output and configured to:determine a first digital value based on the first digital signal;determine a second digital value based on the second digital signal;calculate a resistance ratio based on the first digital value and thesecond digital value; and determine a temperature output value based onthe resistance ratio and the resistor-characterization information. 2.The temperature measurement system of claim 1, wherein: the ADC isfurther configured to: receive a first complementary current signalbased on the resistor; convert the first complementary current signalinto a first complementary digital signal; receive a secondcomplementary current signal based on the thermistor; and convert thesecond complementary current signal into a second complementary digitalsignal; the first digital value is further based on the firstcomplementary digital signal; and the second digital value is furtherbased on the second complementary digital signal.
 3. The temperaturemeasurement system of claim 1, wherein the resistor is located on thesame semiconductor chip as the ADC.
 4. The temperature measurementsystem of claim 1, wherein the resistor-characterization informationcomprises a resistor-characterization ratio based on an actualresistance of the resistor and a designed resistance of the resistor. 5.The temperature measurement system of claim 1, further comprising: alook-up map including a plurality of temperature output values and acorresponding first plurality of resistance-ratio values based on adesigned resistance of the resistor; and a resistance-ratio converterconfigured to calculate a second plurality of resistance-ratio valuesbased on the first plurality of resistance-ratio values and theresistor-characterization information.
 6. A temperature measurementsystem, comprising: a resistor; a resistance-to-current converterconfigured to generate a current signal based on a resistance; ananalog-to-digital converter (ADC) configured to: receive a first currentsignal, the first current signal based on the resistor; convert thefirst current signal into a first digital signal; receive a secondcurrent signal; and convert the second current signal into a seconddigital signal; a memory comprising resistor-characterizationinformation; and a calculation stage communicatively coupled to an ADCoutput and configured to: determine a first digital value based on thefirst digital signal; determine a second digital value based on thesecond digital signal; calculate a resistance ratio based on the firstdigital value and the second digital value; and determine a temperatureoutput value based on the resistance ratio and theresistor-characterization information.
 7. The temperature measurementsystem of claim 6, wherein: the ADC is further configured to: receive afirst complementary current signal, the first complementary currentsignal based on the resistor; convert the first complementary currentsignal into a first complementary digital signal; receive a secondcomplementary current signal; and convert the second complementarycurrent signal into a second complementary digital signal; the firstdigital value is further based on the first complementary digitalsignal; and the second digital value is further based on the secondcomplementary digital signal.
 8. The temperature measurement system ofclaim 6, wherein the resistor is located on the same semiconductor chipas the ADC.
 9. The temperature measurement system of claim 6, whereinthe resistor-characterization information comprises aresistor-characterization ratio based on an actual resistance of theresistor and a designed resistance of the resistor.
 10. The temperaturemeasurement system of claim 6, further comprising: a look-up mapincluding a plurality of temperature output values and a correspondingfirst plurality of resistance-ratio values based on a designedresistance of the resistor; a resistance-ratio converter configured tocalculate a second plurality of resistance-ratio values based on thefirst plurality of resistance-ratio values and theresistor-characterization information.
 11. A method for characterizing atemperature measurement system, comprising: generating a first currentsignal based on a reference resistor; converting the first currentsignal into a first digital signal; determining a first digital valuecorresponding to a resistance of the reference resistor based on thefirst digital signal; generating a second current signal based on a testresistor; converting the second current signal into a second digitalsignal; determining a second digital value corresponding to a resistanceof the test resistor based on the second digital signal; and storingreference-resistor characterization information in a memory.
 12. Themethod of claim 11, further comprising calculating aresistor-characterization ratio based on the first digital value and thesecond digital value, wherein the reference-resistor characterizationinformation comprises the resistor-characterization ratio.
 13. Themethod of claim 11, further comprising: generating a first complementarycurrent signal based on the reference resistor; converting the firstcomplementary current signal into a first complementary digital signal;generating a second complementary current signal based on the testresistor; and converting the second complementary current signal into asecond complementary digital signal; wherein the first digital value isfurther based on the first complementary digital signal; and wherein thesecond digital value is further based on the second complementarydigital signal.
 14. The method of claim 11, wherein the resistance ofthe test resistor is approximately equivalent to a designed resistanceof the reference resistor.
 15. A method for measuring temperature,comprising: generating a first current signal based on a resistor;converting the first current signal into a first digital signal;determining a first digital value corresponding to a resistance of theresistor based on the first digital signal; generating a second currentsignal based on a thermistor; converting the second current signal intoa second digital signal; determining a second digital valuecorresponding to a resistance of the thermistor based on the seconddigital signal; calculating a resistance ratio based on the firstdigital value and the second digital value; and determining atemperature output value based on the resistance ratio and aresistor-characterization ratio.
 16. The method of claim 15, furthercomprising: generating a first complementary current signal based on theresistor, the first complementary current signal complementing the firstcurrent signal; converting the first complementary current signal into afirst complementary digital signal; generating a second complementarycurrent signal based on the thermistor, the second complementary currentsignal complementing the second current signal; and converting thesecond complementary current signal into a second complementary digitalsignal; wherein the first digital value is further based on the firstcomplementary digital signal; and wherein the second digital value isfurther based on the second complementary digital signal.
 17. The methodof claim 15, wherein: converting the first current signal into the firstdigital signal comprises converting the first current signal into afirst stream of digital bits with a continuous-time sigma-deltaanalog-to-digital converter; and converting the second current signalinto the second digital signal comprises converting the second currentsignal into a second stream of digital bits with the continuous-timesigma-delta analog-to-digital converter.
 18. The method of claim 15,wherein calculating the resistance ratio based on the first digitalvalue and the second digital value comprises dividing the first digitalvalue by the sum of the first digital value and the second digitalvalue.
 19. The method of claim 15, wherein calculating the resistanceratio based on the first digital value and the second digital valuecomprises dividing the second digital value by the sum of the firstdigital value and the second digital value.
 20. The method of claim 15,further comprising: calculating a second plurality of resistance-ratiovalues based on a first plurality of resistance-ratio values and theresistor-characterization ratio; and determining which one of the secondplurality of resistance-ratio values is closest to the resistance ratio;and determining a temperature value associated with the closest one ofthe second plurality of resistance ratio values.